Simple and robust digital code tracking loop for wireless communication systems

ABSTRACT

A simple and robust CTL is used for time tracking of multipath components of a spread spectrum signal transmitted over a wireless multipath fading channel. A digital code-tracking loop includes despreading early and late data samples by use of a pseudonoise sequence, an error signal output generated by the despreading, and adjustment for a plurality of on-time, early and late samples, a data rate of a control signal provided as a fractional proportion of a data rate of error signals.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.12/775,969, filed May 7, 2010, which is a continuation of U.S. patentapplication Ser. No. 12/119,139, filed May 12, 2008, which issued asU.S. Pat. No. 7,715,463 on May 11, 2010, which is a continuation of U.S.patent application Ser. No. 10/425,176, filed Apr. 28, 2003, whichissued as U.S. Pat. No. 7,372,892 on May 13, 2008, which claims thebenefit of U.S. Provisional Application No. 60/376,465, filed Apr. 29,2002. All of the above referenced applications are incorporated byreference as if fully set forth.

FIELD OF INVENTION

The present invention relates to the field of wireless communications.More specifically, the present invention relates to an improved codetracking system and method for the field of spread spectrumcommunication systems.

BACKGROUND

Code division multiple access (CDMA) technology has been widely used inmobile cellular phone systems. One of the advantages of CDMA technologyis that it is very robust in scenarios where multiple-path fading may beexperienced. A rake receiver, which is commonly used for CDMA reception,consists of a bank of correlators and a combiner. Each correlator, orrake finger, is used to separately detect and demodulate one of thestrongest multipath components (fingers) of the wideband fading channeland the combiner combines all correlator outputs to obtain the combinedenergy from these strongest multipath components. Since the number ofthe multipath signals and their positions vary in time, time tracking ofeach multipath component is required. For this timing tracking, acode-tracking loop (CTL), also called delay lock loop (DLL), is usuallyused. In previous CTL designs, either a voltage controlled oscillator(VCO) or a numerically controlled oscillator (NCO) was used. A CTL maybe either coherent or noncoherent. Coherent and non-coherent relate tohow the despread data is summed to generate an error signal.

SUMMARY

Apparatus and methods are provided for time tracking of multipathcomponents of a spread spectrum signal transmitted over a wirelessmultipath fading channel. Preferably, a simple and robust code-trackingloop (CTL) includes despreading early and late data samples using apseudonoise sequence, outputting an error signal by the despreading,adjusting for a plurality of on-time, early and late samples, anddetermining a data rate of a control signal as a fractional proportionof a data rate of error signals. The CTL has a simple structure toimplement. A joint CTL is also disclosed for canceling interferencebetween two multipaths when two multipaths are very close to each other.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a diagram of a wireless communications link.

FIG. 2 is a block diagram of CTL using high sampling input data.

FIG. 3 is a block diagram of CTL using low sampling rate input data.

FIG. 4 is a block diagram of one CTL design for UMTS FDD system.

FIG. 5 is a graph showing simulated timing tracking at signal to noiseratio SNR=−24 dB.

FIG. 6 is a graph showing simulated timing tracking at SNR=−24 dB.

FIG. 7 is a graph showing simulated timing tracking at SNR=−24 dB.

FIG. 8 is a graph showing the interference between two adjacent CTLswhen they are separated by less than one and half chip.

FIG. 9 is a block diagram of joint CTL scheme.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

The present invention will be described with reference to the drawingfigures wherein like numerals represent like elements throughout.

FIG. 1 is a diagram of a wireless communications link, which includesone or more base stations 11 (only one shown for simplicity) and one ormore wireless transmit and receive units (WTRUs) 12 (only one shown forsimplicity). The base station includes a transmitter (not shown) andreceiver 13, and the WTRU 12 includes a transmitter (not shown) andreceiver 14. At least one of the base stations 11 and WTRU 12 havetransmit functions so that a communications link is established betweenthe base station 11 and the WTRU 12, as represented by antennas 17, 18.It should be understood by those skilled in the art that the CTL 21 ofthe present invention is implemented within a receiver, such as receiver13 or 14.

A CTL uses the early and late signals (i.e. samples) to generate anerror signal for timing tracking. The early and late samples are definedas the samples that are a half chip (half chip interval) earlier and ahalf chip (half chip interval) later than the on-time sample,respectively. A “chip” is a time interval to transmit one bit ofspreading code and a half chip is half the time interval of a chipinterval. The frequency of a chip time interval is called the “chiprate.” In UMTS CDMA and CDMA2000 standards, the chip rate is defined as3.84 MHz/s.

Referring to FIG. 2, a block diagram of a CTL 21 in accordance with thepresent invention is shown. The inputs are data samples with thesampling rate of 16 times the chip rate. It should be noted thatalthough specific data rates are set forth herein, these data rates areprovided by way of example only. For example, although data sample ratesmay vary, sampling rates of 8 and 16 are typical sample rates. Inanother example using 16 times the rate of sampling, for every 16samples one of the samples will be an “on-time” synchronized samplewhich will be used for despreading, demodulation and rake combining. TheCTL 21 will track this timing and select the on-time sample. To achievethis goal, the CTL 21 will use early and late samples.

CTL 21 includes an input sample selector 23, an early sample pseudonoise(PN) despreader 25, a late sample PN despreader 26, an early-latedetector 27, an integration and dump circuit 28, a sign calculator 29and a summer 30. The input sample selector 23 provides early and latesamples to the PN despreaders 25, 26 which, in turn, provide signals tothe early-late detector 27. The early-late detector 27 includes a latepower calculator 27 a, an early power calculator 27 b and a summer 27 c.The output of the early-late detector 27 is an error signal which isprovided to the integrator and dump circuit 28. The output of theintegrator and dump circuit 28 is sent to the sign calculator 29. Thesign calculator 29 outputs a ±1 signal that is input to the summer 30.The summer 30 converts the relative timing control signal (i.e. −1/+1)to an absolute timing control signal taking into account previousresults. The output of the summer 30 is sent to the input sampleselector 23 to form the loop.

The integration function that is performed by the integrator in theintegration and dump circuit 28 accumulates the signal power and toimprove the signal-to-noise ratio. After the signal is integrated for adefined or predetermined period of time, the integration value isoutput. In order to integrate the signal for the next time period, thesignal in the integrator is first cleared. Accordingly, the procedure inwhich the integrator integrates signal discontinuously between differentperiods of time is called “integration and dump.” The integrationinterval is selected to be a pilot symbol interval. In a preferredembodiment, the pilot symbol interval is a predetermined number ofchips, which in the exemplary embodiment is 256 chips.

The CTL 21 operates by first despreading the early samples and the latesamples. The early and late samples are despread by a PN sequence thatis known to the receiver. The despread data is denoted as S_(e)(k) andS_(l)(k) for early and late samples respectively, where S_(e)(k) andS_(l)(k) are complex numbers, and k represents kth data in the timedomain. The early-late detector 27 uses despread data, or data symbols,to generate an error signal, which can be obtained noncoherently usingEquation (1):

E _(r)(k)=|S _(e)(k)|² −|S _(l)(k)|².  Equation (1)

For each N error signals E_(r)(k), where (N>1), a control signal C₀ willbe generated according to the sign of the sum of these N error signalsE_(r)(k), which can be expressed as:

$\begin{matrix}{C_{o} = {{sign}{\left\{ {\sum\limits_{k = 1}^{N}\; {E_{r}(k)}} \right\}.}}} & {{Equation}\mspace{14mu} (2)}\end{matrix}$

This control signal C₀ is used to adjust all on-time, early and latesamples either forward or backward by M samples. Typically theadjustment is M=1 or 2, or M/16 chip, which is typically 1/16 chip or ⅛chip. The data rate of this control signal C₀ is therefore N times lowerthan the data rate of error signals E_(r)(k).

Still referring to FIG. 2, in some instances the transmitted data can beestimated. If this is the case (i.e., the transmitted data can beestimated), this is done by first removing the modulated data is fromthe despread early signal and despread late signal. This results in:

S _(e)(k)*a(k)*  Equation (3)

and

S _(l)(k)*a(k)*,   Equation (4)

-   -   respectively,        where a(k) is the transmitted symbol or an estimate of        transmitted signal, and ( )* represents the conjugate.        Thereafter, N₁ despread early and late signals with data removed        are coherently summed to calculate the error signal Er(k) that        can be expressed by:

$\begin{matrix}{{E_{r}(k)} = {{{\sum\limits_{k = 1}^{N_{1}}\; {{S_{e}(k)}{a(k)}^{*}}}}^{2} - {{{\sum\limits_{k = 1}^{N_{1}}\; {{S_{l}(k)}{a(k)}^{*}}}}^{2}.}}} & {{Equation}\mspace{14mu} (5)}\end{matrix}$

The despread data S_(e)(k) or S_(l)(k) contains a demodulating symbola(k) that is {−1,+1} for BPSK modulation or {−1,+1,−j,+j} for QPSKmodulation. When the despread data S_(e)(k) or S_(l)(k) is multipliedwith the conjugate of a(k) as in Equations (3) and (4), the a(k)component in the despread data S_(e)(k) or S_(l)(k) will be “removed.”

The data rate of the error signal E_(r)(k) is N₁ times lower than thatof the despread early or late signal since every N₁ despread early orlate signal generates one error signal. For every N error signalsE_(r)(k), where N>1, a control signal C₀ is generated according to thesign of the sum of these N error signals, and the data rate of thiscontrol signal C₀ is N₁×N times lower than the data rate of errorsignals.

In either case, the error signal E_(r)(k) is generated. Equation (1)uses one despread data symbol to generate one error signal E_(r)(k).Equation (5) uses N₁ despread data symbols to generate one error signalE_(r)(k). Therefore the data rates of the error signals E_(r)(k) aredifferent by N₁ times.

According to one embodiment of the present invention, both coherent andnon-coherent approaches are used. Coherent detection adds signalscoherently (i.e. sum the complex numbers directly) such as the sums inEquation 5 (or as will be explained in detail hereafter, the inner sumin Equation 7). Noncoherent detection adds signals noncoherently (i.e.sum the power of complex numbers) such as the sum which will beexplained with reference to Equation 6. The difference between the twoapproaches is that coherent detection has better performance thannoncoherent detection. However, in order to use coherent detection toobtain better performance, the transmitted signal a(k) has to be knownor estimated as performed in Equation 5.

A second embodiment of a CTL 31 in accordance with the present inventionusing low sampling rate input data is shown in FIG. 3. This CTL 31includes an interpolator 33, an early sample PN despreader 35, a latesample PN despreader 36, an early-late detector 37, an integration anddump circuit 38, a sign calculator 39, and a summer 40. The interpolator33 provides early and late samples to the PN despreaders 35, 36, whichin turn provide signals to the early-late detector 37. The early-latedetector 37 includes a late signal power calculator 37 a, an earlysignal power calculator 37 b, and a summer 37 c. The output of theearly-late detector 37 is an error signal E_(r)(k) which is provided tothe integrator and dump circuit 38. The output of the integrator anddump circuit 38 is sent to the sign calculator 39.

The sign calculator 39 outputs a ±1 signal that is supplied to thesummer 40. The summer 40 converts the relative timing control signal(i.e. −1/+1) to an absolute timing control signal taking into accountprevious results. The output of the summer 40 is sent to theinterpolator 33 to form the loop in the same manner as depicted in FIG.2.

For low sampling rate input data, the sampling rate is typically twosamples per chip. In order to adjust the timing for on-time andearly/late samples forward or backward by a fraction of chip (forexample 1/16 chip or ⅛ chip), the interpolator 33 is used to generateall on-time samples, and early/late samples which are offset by suchamount of time from the previous samples.

As can be seen, the input data rates are different for the input sampleselector 23 shown in FIG. 2 and the interpolator 33 shown in FIG. 3. Thesample selector 23 selects which input samples to use according to thecontrol signal C₀. Since the interpolator 33 has only two input samplesper chip, it has to regenerate or interpolate the desired samplesaccording to a control signal input.

The CTL 21 of FIG. 2 requires a high-speed analog-to-digital converter(ADC). The CTL 31 of FIG. 3 uses a low-speed ADC, which is lower incost, but CTL 31 also requires an extra interpolator to regenerate thedesired samples. With CTL 21, a high data rate (i.e. 16 samples/chip) isused and therefore a high speed ADC is required. With CTL 31, a low datarate (i.e. 2 samples/chip) is used and therefore a low speed ADC isrequired. The different data rates are needed for differentapplications. For example, in FIG. 4, a low speed ADC is preferredbecause is uses 2 samples/chip and interpolator 53.

In an exemplary embodiment corresponding to the UMTS FDD standard, foruplink transmissions every slot of the dedicated physical controlchannel contains ten symbols (including pilot, transmit power controland TFCI bits). Among these ten symbols, pilot symbols are known to thereceiver, but the power control and TFCI bits are unknown to thereceiver. Suppose that SE_(k,j) and SL_(k,j) denote the despread earlyand late signals for the jth symbol in the kth slot. If the CTL 31 isupdated every two frames (there are 15 slots per frame and 30 slots pertwo frames), then the control signal C₀ at the output of the integrationand dump circuit 38 using noncoherent combining can be expressed as:

$\begin{matrix}{C_{o} = {{SIGN}{\left\{ {\sum\limits_{k = 1}^{30}\; {\sum\limits_{j = 1}^{10}\; \left\{ {{{SE}_{k,j}}^{2} - {{SL}_{k,j}}^{2}} \right\}}} \right\}.}}} & {{Equation}\mspace{14mu} (6)}\end{matrix}$

Alternatively CTL 31 coherently sums a number of early and late signalsfrom one slot, and then calculates the power and the error signalE_(r)(k). Again if the CTL 31 is updated every two frames, then thecontrol signal C0 at the integrator output can be expressed as:

$\begin{matrix}{{C_{o} = {{SIGN}\left\{ {\sum\limits_{k = 1}^{30}\; \left\{ {{{\sum\limits_{j = 1}^{N_{1}}\; {{SE}_{k,j}a_{k,j}^{*}}}}^{2} - {{\sum\limits_{j = 1}^{N_{1}}\; {{SL}_{k,j}a_{k,j}^{*}}}}^{2}} \right\}} \right\}}};} & {{Equation}\mspace{14mu} (7)}\end{matrix}$

where a_(k,j) is the known pilot bit or the estimated power control/TFCIbit in the jth symbol of the kth slot.

Some further alternatives are possible by implementing variouscombinations of the following items: 1) using an input sample selector23 (for the high speed ADC as shown in FIG. 2) or interpolator 33 (forthe low speed ADC as shown in FIG. 3); 2) using a noncoherent errorsignal calculation as in Equations 1 and 6 or using coherent errorsignal calculation as in Equations 5 and 7; and 3) using error signalpower as in Equations 1-5, 6 and 7 or using error signal absolute valueas in Equation 9. As explained above, FIG. 2 uses an input sampleselector, noncoherent error signal calculation, and error signal power(Equation 1) and FIG. 3 uses an interpolator, noncoherent error signalcalculation and error signal power (Equation 1). FIG. 4, explainedbelow, uses an interpolator, noncoherent error signal calculation anderror signal absolute value.

As explained above Equations (6) and (7) represent two different methodsto generate the error signal E_(r)(k) as explained above. Equation (6)uses noncoherent detection and uses the error signal generation inEquation (1), and Equation (7) uses coherent detection and uses theerror signal generation in Equation (5). The “SIGN” is used to adjustthe timing forward or backward. When the sign of Equations (6) or (7) ispositive, it will adjust the timing backward; whereas when the sign ofEquations (6) or (7) is negative, it will adjust the timing forward.

An embodiment of a CTL for UMTS FDD in accordance with the presentinvention is shown in FIG. 4. The CTL circuit 51 includes aninterpolator 53, a delay circuit 54, early and late PN despreaders 55,56, two magnitude calculation circuits 57, 58 which calculate absolutevalues of the respective signals, and a summer 59. Also included is anintegrator and dump circuit 63, a sign calculator 64 and a second summer65. The interpolator 53 provides a single early/late output to delaycircuit 54, which provides an early signal to early PN despreader 55.The output of interpolator 53 is provided directly to late PN despreader56 and the outputs of the despreaders 55, 56 are provided to respectivemagnitude calculation circuits 57, 58.

The circuit of FIG. 4 uses the first error signal generation methoddescribed by Equations (1) and (6) because the early sample and latesample are separated by exactly one chip interval, and the early samplecan be obtained from late sample by delaying one sample. Further, inFIG. 4, the square calculation performed by the early and late signalpower calculators 37 a, 37 b is replaced with an absolute valuecalculation in order to simplify the hardware complexity.

If one compares Equation (9) with Equation (1), it will be noted thatthe integrator and dump circuit 63 performs the summing as described inEquation (6); and the sign calculator 64 resolves the sign (+or −) asdescribed in Equation (6). Since this sign generates a relative timingadjustment, a new absolute timing signal is generated by summing theprevious absolute timing with the incoming relative adjustment. This isdone in summer 65.

The absolute values (of the early and late despreaders 55, 56 calculatedin the magnitude calculation circuits 57, 58) are provided to the summer59, which provides an error signal E_(r)(k) as its output to theintegrator and dump circuit 63 which and, in turn, outputs to the signcalculator 64. The output from the sign calculator 64 hard limited to a±1 signal, which is supplied as a phase control signal to theinterpolator 53, to form the loop.

The error signal Δ_(k,j) is the difference of the absolute values ofE_(k,j) and L_(k,j), which can be expressed as:

Δ_(k,j) =|E _(k,j) |−|L _(k,j)|  Equation (9)

The integrator and dump circuit 63 provides the magnitude of the errorsignals and its output is hard-limited by the sign calculator 64 toeither +1 or −1 according to the sign of the summed error signals. This+1 or −1 is used to adjust the timings of all punctual, early and latesamples by ⅛ chip forward or backward and is implemented by controllingthe interpolator phase. This interpolator phase is updated bysubtracting the previous phase with the new input data (+1 or −1).

The interpolator 53 uses four samples (with the sampling interval of ahalf chip) to generate the punctual and late samples. The relationshipbetween the phase control signal (i.e. the interpolator output), thetiming offset and the interpolator coefficients is shown in Table 1. Theearly sample is generated by delaying one sample of the previouslygenerated late sample. If the punctual sample is on phase “0,” then thelate sample will be on the phase “2.” If the punctual sample is on phase“x,” then the late sample will be on phase “x+2.”

TABLE 1 Interpolator Phase, Timing Offset and Coefficients. timingInterpolator offset coefficient coefficient coefficient coefficientPhase (chips) 1 2 3 4 −6 −0.7500 0.0000 0.0000 0.0000 1.0000 −5 −0.62500.0547 −0.2578 0.6016 0.6016 −4 −0.5000 0.0625 −0.3125 0.9375 0.3125 −3−0.3750 0.0391 −0.2109 1.0547 0.1172 −2 −0.2500 0.0000 0.0000 1.00000.0000 −1 −0.1250 −0.0391 0.2734 0.8203 −0.0547 0 0.0000 −0.0625 0.56250.5625 −0.0625 1 0.1250 −0.0547 0.8203 0.2734 −0.0391 2 0.2500 0.00001.0000 0.0000 0.0000 3 0.3750 0.1172 1.0547 −0.2109 0.0391 4 0.50000.3125 0.9375 −0.3125 0.0625 5 0.6250 0.6016 0.6016 −0.2578 0.0547 60.7500 1.0000 0.0000 0.0000 0.0000

The integrator and dump 63 is reset every 30 slots during steadytracking mode, and is reset every ten slots during the initial pull-inmode. At the beginning, the CTL 51 is in a “rough” timing position. Itis desirable for CTL 51 to react quickly to find the right timingposition (initial pull-in mode), and then the CTL 51 will lock to thisposition and track any timing change (tracking mode). During the firstfive frames after the finger is assigned to the CTL 51, the CTL 51 isassumed to be in the pull-in mode, and from the sixth frame on, the CTL51 is assumed to be in the tracking mode.

For the pull-in mode, the CTL 51 is updated every ten slots and all tenpilot and data symbols are used per dedicated physical control channel(DPCCH) slot. In this case the accumulator output Q can be expressed as:

$\begin{matrix}{Q = {{SIGN}{\left\{ {\sum\limits_{k = 1}^{10}\; {\sum\limits_{j = 1}^{10}\; \Delta_{k,j}}} \right\}.}}} & {{Equation}\mspace{14mu} (10)}\end{matrix}$

For steady mode, the CTL 51 is updated every 30 slots (or two frames)and all ten pilot and data symbols are used per DPCCH slot. Theintegrator and dump circuit output 63 can be expressed as:

$\begin{matrix}{Q^{\prime} = {{SIGN}{\left\{ {\sum\limits_{k = 1}^{30}\; {\sum\limits_{j = 1}^{10}\; \Delta_{k,j}}} \right\}.}}} & {{Equation}\mspace{14mu} (11)}\end{matrix}$

Simulations of the results of CTL 51 tracking during a steady mode wereperformed, The simulation parameters were as follows:

-   1) Both time and frequency drift is 0.613 ppm;-   2) The channel is AWGN channel;-   3) Target SNR=−24 dB;-   4) The CTL 51 is updated every two frames (30 slots);-   5) For each CTL 51 updating, ⅛ chip forward or backward adjustment    is applied;-   6) The maximum timing error is calculated;-   7) The root-square of mean square timing error (RMSE) is calculated;-   8) Both noncoherent and coherent combining are considered;-   9) For noncoherent combining, ten symbols per slot are used, and the    error signal calculation is same as Equation (6);-   10) For coherent combining, only three pilot symbols per slot are    used, and the error signal calculation is same as Equation (7) with    N1=3;-   11) A simplified scheme is simulated, which uses the absolute value    instead of power of early and late signals.

FIG. 5 is a graph showing simulated timing tracking at SNR=−24 dB usingcoherent detection. By applying Equation (7), a noncoherent combining often pilot symbols per slot is achieved. FIG. 6 is a graph showingsimulated timing tracking at SN =−24 dB using non-coherent detection.

FIG. 7 shows the results of a simplified error signal calculation inaccordance with the present invention using Equation (11). Since theerror signal calculation in both Equation (6) for noncoherent combiningand Equation (7) for coherent combining need to calculate the power ofcomplex numbers, this power calculation is very complicated in ahardware implementation. In order to reduce the hardware complexity, themagnitude calculation is used instead of the power calculation.

If all ten pilot and data symbols are used for noncoherent combining ineach slot and the CTL is updated every two frames (30 slots), then theaccumulator output can be expressed as:

$\begin{matrix}{Q^{''} = {{SIGN}{\left\{ {\sum\limits_{k = 1}^{30}\; {\sum\limits_{j = 1}^{10}\; \left\{ {{E_{k,j}} - {L_{k,j}}} \right\}}} \right\}.}}} & {{Equation}\mspace{14mu} (12)}\end{matrix}$

If only first three pilot symbols are used for coherent combining ineach slot and the CTL is updated every two frames (30 slots), then theaccumulator output can be expressed as:

$\begin{matrix}{Q = {{SIGN}{\left\{ {\sum\limits_{k = 1}^{30}\; \left\{ {{{\sum\limits_{j = 1}^{3}\; E_{k,j}}} - {{\sum\limits_{j = 1}^{3}\; L_{k,j}}}} \right\}} \right\}.}}} & {{Equation}\mspace{14mu} (13)}\end{matrix}$

Table 2 is a set of performance comparisons of the RMSE of differenceCTL schemes. In this table, three CTL schemes were compared. One is thenoncoherent combining using ten symbols per slot; the second is thecoherent combing using three pilot symbols per slot; and the third isthe simplified noncoherent combining using ten symbols per slot. For thetarget SNR=24 dB, the three schemes perform closely. When the SNR is −34dB, the coherent combining performs worst because fewer symbols areused. The simplified scheme is worse than the non-simplified version.

TABLE 2 The RMSE of difference CTL schemes Noncoherent CoherentSimplified combining using combining noncoherent 10 symbols using 3symbols combining using per slot per slot 10 symbols per slot SNR = −241.63 1.51 1.52 dB SNR = −30 2.18 2.27 2.17 dB SNR = −34 3.07 5.15 4.03dB

Each CTL tracks one finger independently. When two multipaths (orfingers) are within one and half chip, the two CTLs for the two fingerswill interfere with each other and therefore degrades the CTL trackingperformance. According to a particular aspect of the invention, a jointCTL scheme is used to reduce the interference from each other. Withoutloss of generality, it is possible to take an approach that there aretwo multipaths. The received signal r(t) can be expressed as

r(t)=h ₁(t)s(t)+h ₂(t)s(t−τ)  Equation (14)

where s(t) is the useful signal,

${{s(t)} = {\sum\limits_{k = {- \infty}}^{\infty}\; {a_{k}{g\left( {t - {kT}} \right)}}}},$

a_(k) is the information symbol and g(t) is the signal waveform. h₁ (t)is the channel gain of the first path and h₂(t) is the channel gain ofthe second path. τ is the relative delay between the two fingers. Notethat the additive white Gaussian noise is not considered in Equation(14).

When the relative delay between two adjacent fingers is less than 1.5chip, the two independent CTLs will interfere with each other as shownin FIG. 8. It should be noted that triangle waveform is used fordemonstration only and is not necessarily used in practice. Due to theinterference, the performance of the two CTLs will degrade. The sampleof the late signal of the first finger will contain the interferenceh₂g(τ−T/2) from the second finger, and the sample of the early signal ofthe second finger will contain the interference h₁g(τ−T/2) from thefirst finger. The sample of the late signal of the first finger S_(l)^(1st)(k) is:

S _(l) ^(1st)(k)=h ₁(k)g(T/2)+h ₂(k)g(τ−T/2)  Equation (15)

and the sample of the early signal of the second finger S_(e) ^(2nd)(k)is:

S _(e) ^(2nd)(k)=h ₁(k)g(τ−T/2)+h ₂(k)g(T2).  Equation (16)

FIG. 9 is a block diagram of joint CTL scheme 100. The components aresimilar to FIG. 4, but with a joint error signal calculator 102operating as part of two CTL circuits 103, 104.

CTL circuit 103 includes an interpolator 113, a delay circuit 114, earlyand late PN despreaders 115, 116, to magnitude calculation circuits 117,118 which calculate absolute values of the respective signals, and to asummer 119. Also included is an integrator and dump circuit 123, a signcalculator 124, and a second summer 125. The interpolator 113 provides asingle early/late output to delay circuit 114, which provides an earlysignal to early PN despreader 115. The output of interpolator 113 isprovided directly to late PN despreader 116 and the outputs of thedespreaders 115, 116 are provided to respective magnitude calculationcircuits 117, 118. CTL circuit 104 includes an interpolator 133, a delaycircuit 134, early and late PN despreaders 135, 136, to magnitudecalculation circuits 137, 138 which calculate absolute values of therespective signals, and to a summer 139. Also included is an integratorand dump circuit 143, a sign calculator 144, and a second summer 145.The interpolator 133 provides a single early/late output to delaycircuit 134, which provides an early signal to early PN despreader 135.The output of interpolator 133 is provided directly to late PNdespreader 136 and the outputs of the despreaders 135, 136 are providedto respective magnitude calculation circuits 137, 138.

As can be seen, the relative delay τ between the two fingers can beobtained from two CTLs. As is the case with the circuit of FIG. 4, thecircuit of FIG. 9 uses the first error signal generationmethod-described by Equations (1) and (6) because the early sample andlate sample are separated by exactly one chip interval and the earlysample can be obtained from late sample by delaying one sample. Anabsolute calculation is used in order to simplify the hardwarecomplexity.

According to particular aspects of the present invention, the followingtwo methods are effective to cancel interference:

Method 1: If the channel gains h₁(t) and h₂(t) , are known, theinference is cancelled by subtracting the interference from usefulsignal. The error signals are generated as

E _(r) ^(1st)(k)=|S _(e) ^(1st)(k)|² −S _(l) ^(1st)(k)−h₂(k)g(τ−T/2)|²  Equation (17)

E _(r) ^(2nd)(k)=|S _(e) ^(2nd)(k)−h ₁(k)g(τ−T/2)|²−|S _(l)^(2nd)(k)|²  Equation (18)

The control signal Co is calculated using Equation (2).

Method 2: If the channel gains h₁ and h₂ are not known, but the power ofthe two fingers is known, which are the means of the channel gains |h₁|²and |h₂|², E|h₁|² and E|h₂|². Since:

$\begin{matrix}{{\frac{1}{N}{\sum\limits_{k = 1}^{N}\; {{S_{l}^{1\; {st}}(k)}}^{2}}} = {{E{h_{1}}^{2}{g^{2}\left( {T/2} \right)}} + {E{h_{2}}^{2}{g^{2}\left( {\tau - {T/2}} \right)}}}} & {{Equation}\mspace{14mu} (19)} \\{{\frac{1}{N}{\sum\limits_{k = 1}^{N}\; {{S_{l}^{2\; {nd}}(k)}}^{2}}} = {{E{h_{1}}^{2}{g^{2}\left( {\tau - {T/2}} \right)}} + {E{h_{2}}^{2}{g^{2}\left( {T/2} \right)}}}} & {{Equation}\mspace{14mu} (20)}\end{matrix}$

The control signal C₀ is calculated as follows with the interferenceremoved.

$\begin{matrix}{C_{0}^{1\; {st}} = {{sign}\left\{ {{\frac{1}{N}{\sum\limits_{k = 1}^{N}\; {{S_{e}^{1\; {st}}(k)}}^{2}}} - {\frac{1}{N}{\sum\limits_{k = 1}^{N}\; {{S_{l}^{1\; {st}}(k)}}^{2}}} - {E{h_{2}}^{2}{g^{2}\left( {\tau - {T/2}} \right)}}} \right\}}} & {{Equation}\mspace{14mu} (21)} \\{C_{0}^{2\; {nd}} = {{sign}\left\{ {{\frac{1}{N}{\sum\limits_{k = 1}^{N}\; {{S_{e}^{2\; {nd}}(k)}}^{2}}} - {\frac{1}{N}{\sum\limits_{k = 1}^{N}\; {{S_{l}^{2\; {nd}}(k)}}^{2}}} - {E{h_{1}}^{2}{g^{2}\left( {\tau - {T/2}} \right)}}} \right\}}} & {{Equation}\mspace{14mu} (22)}\end{matrix}$

The present invention is useful in cellular mobile systems. In onepreferred embodiment, the invention is implemented in a base stationtransmission as controlled by a radio network controller or a Node Btransmit controller. It is understood, however, that the invention canbe used for a wide variety of spread spectrum communicationstransmissions.

1. A digital code-tracking loop comprising: a despreader for despreadingearly and late data samples by use of a pseudonoise sequence; an errorsignal output generated by the despreader; and an adjustment for aplurality of on-time, early and late samples, a data rate of a controlsignal provided as a fractional proportion of a data rate of errorsignals.
 2. The digital code-tracking loop of claim 1, wherein: thedespread data samples include S_(e)(k) and S_(l)(k) for early and latesamples, respectively, S_(e)(k) and S_(l)(k) provided as a complexnumber, k represents the kth data in a time domain; and the early andlate data rendersE _(e)(k)=|S _(e)(k)|² −|S _(l)(k)|².
 3. The digital code-tracking loopof claim 1, comprising: the error signal output providing timingtracking, the early and late samples defined as the samples at half chipinterval earlier and half chip interval later than on-time values,respectively.
 4. The digital code tracking loop of claim 3, wherein forevery N samples, one sample provides an on-time synchronized sample,used for despreading, demodulation and rake combining, the code trackingloop tracking this timing and selecting the on-time sample.
 5. Thedigital code tracking loop of claim 3, comprising: each of a pluralityof slots of a dedicated physical control channel include 10 symbols, the10 symbols providing pilot, transmit power control and TFCI bits; andthe code tracking loop updated every 2 frames.
 6. The digitalcode-tracking loop of claim 1, wherein the adjustment provides timetracking of multipath component of direct sequence spread spectrumsignal over a wireless multipath fading channel.
 7. A radio transmissioncontroller implementing a digital code-tracking loop, the radiotransmission controller comprising: a despreading circuit capable ofdespreading early and late data samples by use of a pseudonoisesequence; a circuit providing an error signal output generated by thedespreading; a circuit providing a control signal; and a circuitproviding adjustment for a plurality of on-time, early and late samples,a data rate of a control signal provided as a fractional proportion of adata rate of error signals.
 8. The radio transmission controller ofclaim 7, wherein: radio transmission controller despread data sampleswhich include S_(e)(k) and S_(l)(k) for early and late samples,respectively, S_(e)(k) and S_(l)(k) provided as a complex number, krepresents the kth data in a time domain; and the despreading circuitproviding the early and late data samples according toE _(e)(k)=|S _(e)(k)|² |S _(l)(k)|².
 9. The radio transmissioncontroller of claim 7, comprising a circuit for proving an error signaloutput providing timing tracking , so as to provide the early and latesamples at half chip interval earlier and half chip interval later thanon-time values, respectively.
 10. The radio transmission controller ofclaim 9, wherein for every N samples, one sample provides an on-timesynchronized sample, used for despreading, demodulation and rakecombining, the code tracking loop tracking this timing and selecting theon-time sample.
 11. The radio transmission controller of claim 9,comprising: each of a plurality of slots of a dedicated physical controlchannel include 10 symbols, the 10 symbols providing pilot, transmitpower control and TFCI bits; and the code tracking loop updated every 2frames.
 12. The radio transmission controller of claim 7, wherein theadjustment provides time tracking of multipath component of directsequence spread spectrum signal over a wireless multipath fadingchannel.
 13. The radio transmission controller of claim 7, comprising ajoint error signal calculator circuit provided as the circuit providingthe error signal for the despreading circuit and for the second digitalcode tracking loop.
 14. The radio transmission controller of claim 13,wherein the joint error signal calculator provides an indication of arelative delay τ between two fingers of a composite signal.
 15. Theradio transmission controller of claim 14, wherein the relative delay τbetween the two fingers provides an indication of a delay for signalinterference calculation.